8.2.2.1 Primary Bulk Capacitance

The primary side bulk cap, C4, is selected based on the power level and the desired minimum bulk voltage level. The bulk capacitor value can be calculated as:

Equation 10.

where

• P_IN is the maximum input power. Input power is the maximum output power divided target efficiency.
• V_IN(min) is the minimum AC input voltage RMS value.
• V_BULK(min) is the target minimum bulk voltage.
• f_LINE is the line frequency.

Based on the equation, to achieve 70-V minimum bulk voltage, the bulk capacitor should be larger than 72 µF and 82 µF was chosen in the design.

8.2.2.2 Transformer

The transformer design starts with selecting a suitable switching frequency. Generally, the switching frequency selection is based on the tradeoff between the converter size and efficiency. Higher switching frequency results in smaller transformer size, but the switching losses will increase, potentially impacting efficiency. Sometimes, the switching frequency is selected to avoid certain frequencies or harmonics that could interfere with those used for communication. The frequency selection is beyond the scope of this datasheet.

EMI regulations place limits on EMI noise at 150 kHz and higher. For this design, 145 kHz is selected for the switching frequency to minimize transformer size while keeping the switching frequency below the EMI regulation band.

The transformer turns ratio can be selected based on the desired MOSFET voltage rating and diode voltage rating. Since the maximum input voltage is 130 V AC, the peak bulk voltage can be calculated as:

To take advantage of the low Rdson of lower voltage MOSFETs, a target device rating of 400 V is selected. Considering the design margin and extra voltage ringing on the MOSFET drain, the reflected output voltage should be less than 50 V. The transformer primary to secondary (n_PS) turns ratio can be selected as:

Equation 12.

The output rectifier diode (D4) voltage stress is also affected by the turns ratio. The stress applied to the diode is the output voltage plus the reflected input voltage. The voltage stress on the diode can be calculated as:

Equation 13.

Considering the ringing voltage spikes always present in a switching power supply and allowing for voltage derating (normally 80% derating is used), the diode voltage rating should be higher than 150 V.

The transformer inductance selection is based on the requirement for this converter to remain in discontinuous conduction (DCM). Selecting a larger inductance would allow the converter operate in continuous conduction (CCM). CCM operation tends to increase the transformer size. The primary inductance (L_m) can be calculated as:

Equation 14.

In this equation, f_sw is the 145-kHz switching frequency. Therefore, the transformer inductance should be selected as 85 µH.

The auxiliary winding provides the power for LM5021 during normal operation. The auxiliary winding voltage is the output voltage reflected to the primary side. A higher reflected voltage allows the IC to quickly get energy from the transformer during startup and makes starting a heavy or highly capacitive load easier. However, a high auxiliary reflected voltage makes the IC consume more power, reducing efficiency and increasing standby power consumption. Therefore, a tradeoff is required. In this design, the auxiliary winding voltage is selected to ensure that there is enough voltage available to ensure the controller will operate when the output voltage is programmed to the lower 8-V setting. Therefore, the auxiliary winding to the output winding turns ratio is selected as:

Equation 15.

8.2.2.3 Main Switch FET and Output Rectifier

Based on calculated inductor value and the switching frequency, the current stress of the MOSFET (Q4) and diode (D4) can be calculated.

The peak current of Q4 can be calculated as:

Equation 16.

The peak current is 2.55 A.

The peak current in D4 is the peak current in Q4 reflected to the secondary side:

The RMS current in Q4 can be calculated as:

Equation 18.

Here D is the Q4 on time duty cycle at minimum bulk voltage and it can be calculated as:

Equation 19.

The RMS current in Q4 is 0.97 A. Therefore, STP11NK40ZFP is selected.

The average current in D4 is the output current 1.45 A. With a 150-V reverse voltage rating and a 20-A average current rating, 20CTQ150 is selected.

The output capacitor is selected based on the output voltage ripple requirement. In this design, 0.1% voltage ripple is assumed. Based on the 0.1% ripple requirement, the capacitor value can be selected based on:

Equation 20.

Considering the tolerance and temperature effect, together with the ripple current rating of the capacitors, the output capacitor is selected as two 220 uF units in parallel.

8.2.2.4 Timing Resistor

The switching frequency is set by R17. From Equation 2:

Equation 21.

Choose R17 as 22.1 k as a common resistor close to the computed value.

8.2.2.5 Soft-Start Time

The soft start time is set by C14. This determines the rate of increase of converter primary peak current at startup. Set a time that is long enough so that the feedback lop can compensate for the transition from open loop during soft start to being closed loop as it takes over from soft start. The value is best determined experimentally after the rest of the converter is complete. For this example, 220 nF was chosen as the best fit for startup time and startup transient overshoot.

8.2.2.6 Current Sensing Network

The current sensing network consists of C15, R23, R22 and optionally R21. R23 sets the maximum peak current in the transformer primary. Given a peak current of 2.5 A:

Equation 22.

Select R23 to be 0.2 Ω.

R22 and C15 form a pulse filter that helps provide additional immunity beyond the internal blanking time to the sudden voltage spike produced on R23 by the parasitic capacitance of the transformer and snubber network for Q4. The time constant for this filter is best determined experimentally but as a guideline should be no more than 25% of the minimum pulse width of the converter in actual operating conditions. Keeping the impedance low also helps with preventing noise coupling problems. For this converter 100 Ω and 150 pF were selected to give a time constant of 67 ns.

R21 is used to disable pulse skip mode if that is needed. To disable pulse skip mode, R21 must produce a 125 mVdc level or slightly higher at the CS pin. To calculate the required value:

Equation 23.

Since VCC is 8 V:

Select R21 to be 6.49 k to disable skip mode operation.

8.2.2.6.3 Startup Circuit

The startup circuit for this converter illustrates a technique for starting the converter quickly without the need to wait for the larger VIN capacitance to be trickle charged through high impedance from the bulk voltage. It also allows the steady state impedance connected to the bulk voltage to be higher than otherwise possible, reducing power dissipation. The circuit consists of a series pass regulator (R1, R2, R5, Q1, D5 and C6) from the bulk voltage supply to VIN, a series pass regulator from the rectified AUX winding to VIN (C12, D8, Q3, R12 and D9) and a turn off circuit that turns off the bulk regulator once the converter is running (D7, R10 and Q2).

Q1 is selected for small size and the ability to withstand the maximum bulk voltage. A BSS127 is selected for its high maximum drain voltage of 600 V.

R2 is selected as 10k simply to limit current through Q1 to less than 50 mA per the BSS127 data sheet, at the maximum possible bulk voltage.

Equation 25.

The voltage the bulk regulator supplies is equal to the zener voltage of D5 less the threshold voltage of Q1, typically 4 V. Since the controller requires a maximum of 23 V to start, the zener voltage must be at least 27 V. D5 is selected as a 27 V device, BZX585-C27.

The bulk series regulator is turned on by R1 and R5. These are only required to supply enough current to bias D5 and overcome any leakage in Q2, approximately 10 µA. To guarantee operation, bias the circuit with 25 µA minimum.

Equation 26.

The sum of R1 and R2 must be less than 8.6 MΩ. R1 and R2 are selected as 1.5 M each for common values.

C6 is simply a time delay to soften the startup of the bulk regulator and was arbitrarily chosen as 10 nF.

Turning to the AUX regulator, D8 protects the b-e junction of Q3 while the bulk regulator is active. It is a low current device that must withstand 27 V minimum. A TS4148 was chosen for this purpose.

Q3 is the pass element for the AUX regulator and is again low current. The only requirement is that the Vce rating be greater than the maximum rectified AUX voltage of 36 V. A common MMBT2222A was chose for this function.

The voltage supplied to the controller VIN pin by the AUX regulator is determined by voltage on C13 less 1.4 V for the drop across D8 and Q3 when the voltage on C13 is less than the zener voltage of D9 or the zener voltage of D9 less 1.4 V when the voltage on C13 is higher than the zener voltage of D9. The maximum voltage supplied to the controller is the zener voltage of D9 less 1.4 V. Picking a zener voltage of 15 V lets the controller run at approximately 13.6 V under normal conditions. A BZX585-C15 is selected.

R12 provides bias for D9 and base drive for Q3. Bias for D9 is small compared to the required base drive for Q3. The current required from the regulator is the sum of the controller operating current and the drive current for Q4. The drive current for Q4 depends on the operating frequency and the total gate charge of Q4 at 13.6 V.

Equation 27. I_REG > I_VIN + I_DRV

Equation 28. I_VIN = 3.5 mA

Equation 29. I_DRV = 40 nC × 145 kHz = 5.8 mA

The regulator must supply a minimum of about 9.5 mA for the controller to function. From the MMBT2222A datasheet, the minimum current gain is 75 for 10-mA collector current. The worst case base drive occurs when the output is programmed for 8 V, giving 12 V available to the collector of Q3 and to the base drive resistor R12. To get a minimum of 10 mA from the regulator requires 133 µA of base drive current. The VIN voltage cannot fall below 7.25 V or the controller will shut down. R12 must satisfy the following relationship:

Equation 30.

R12 was picked as 22 k for this application.

The bulk regulator turn off circuit simply turns Q1 off when the voltage on C13 exceeds the zener voltage of D7 plus the b-e voltage of Q2 (0.7 V) and whatever voltage is required to get sufficient base drive through R10. The required collector current is at the maximum bulk voltage of 255 V.

Equation 31.

Current gain for the selected MMBT2222A is over 100 at this level so the base drive required is only 8.5 µA. Picking R10 as 47 k requires only an additional 400 mV on C13 to effect turn off of the bulk regulator. The bulk regulator should turn off before the voltage on C13 reaches 12 V. Picking a zener voltage of 10 V with the BZX585-C10 ensures that the bulk regulator will turn off at no more than 11.8 V.