8.1.1 Switching Frequency

Switching frequency is selected based on the trade-offs between efficiency, solution size/cost and the range of output voltage that can be regulated. Many applications place limits on switching frequency due to EMI sensitiviy. The on-time of the TPS92640 and TPS92641 devices can be programmed for switching frequencies ranging from the tens of kHz to over 1 MHz. This on-time varies in proportion to both V_IN and V_OUT, as described in Switching Frequency. However, in practice the switching frequency will shift in response to large swings in input or output voltage. The maximum switching frequency is limited only by the minimum on-time and minimum off-time requirements.

8.1.2 LED Ripple Current

The LED manufacturers generally recommend values of current ripple, ΔI_LED, to achieve optimal optical efficiency. The peak-to-peak current ripple values typically range from ±10% to ±40% of DC current, I_LED. Higher LED ripple current allows the use of smaller inductors, smaller output capacitors, or no output capacitors at all. Lower ripple current requires more inductance, higher switching frequency, or additional output capacitance. Based on the LED current ripple specification and desired switching frequency, the inductor value can be calculated using Equation 13.

Equation 13.

It is important to ensure that the rated inductor saturation current is greater than the worst case operating current (I_LED+ΔI_LED/2) under the wide operating temperature range.

8.1.3 Buck Converters Without Output Capacitor

A Buck regulator is ideal for regulating current because of the direct connection between the inductor and the LED load. Because the current is being regulated, not voltage, a buck current regulator is free of load current transients, and has no need of output capacitance to supply the load and maintain output voltage. This is of great benefit when driving LEDs as large electrolytic capacitors impact the lifetimes and PWM dimming performance. The output capacitor can be eliminated by using a large inductor or higher switching frequency as discussed in LED Ripple Current

A capacitor placed in parallel with the LED or array of LEDs can be used to reduce Δi_LED while keeping the same average current through both the inductor and the LED array. With this topology the inductance can be lowered, making the magnetics smaller and less expensive. Alternatively, the circuit can be run at lower frequency with the same inductor value, improving the efficiency and expanding the range of output voltage that can be regulated.

Figure 22 shows the equivalent impedances presented to the Δi_L-PP when an output capacitor, C_OUT, and its equivalent series resistance (R_ESR) are placed in parallel with the LED array.

Figure 22. LED Ripple Current With C_OUT

To calculate the respective ripple currents, the LED array is represented as the dynamic resistance, (r_D). LED's dynamic resistance is not always specified on the manufacturer’s data sheet, but it can be calculated as the inverse slope of the LED’s V_LED vs I_LED curve at the operating point. However, this method only gives an rough estimate of r_D. Total dynamic resistance for a string of n LEDs connected in series can be calculated as the r_D of one device multiplied by n. Inductor ripple current, Δi_L-PP is still calculated as before. The following equations can then be used to estimate peak-to-peak LED current ripple, Δi_LED-PP, when using a parallel capacitor:

Equation 14.

The calculation for Z_COUT assumes that the shape of the inductor ripple current is approximately sinusoidal. Small values of C_OUT that do not significantly reduce Δi_LED-PP can also be used to control EMI generated by the switching action of the TPS92640 and TPS92641 devices. EMI reduction becomes more important as the length of the connections between the LED and the rest of the circuit increase.

8.1.4 Input Capacitor

Input capacitor is selected using requirements for minimum capacitance and rms ripple current. The input capacitor supply pulses of current approximately equal to I_LED while the high-side NFET is on, and is charged up by the input voltage while the high-side NFET is off. Switching converters such as the TPS92640 and TPS92641 devices have a negative input impedance due to the decrease in input current as input voltage increases. This inverse proportionality of input current to input voltage can cause oscillations (sometimes called power supply interaction) if the magnitude of the negative input impedance is greater than the input filter impedance. Minimum capacitance can be selected by comparing the input impedance to the converter’s negative resistance; however, this requires accurate calculation of the input voltage source inductance and resistance, quantities which can be difficult to determine. An alternative method to select the minimum input capacitance (C_IN-MIN) is to select the maximum voltage ripple (Δv_IN-MAX), which can be tolerated. Δv_IN-MAX is equal to the change in voltage across C_IN during t_ON when it supplies the load current. A good starting point for selection of C_IN is to use an input voltage ripple of 2% to 10% of V_IN. C_IN-MIN can be selected using Equation 15.

Equation 15.

TI recommends a minimum input capacitance at least 75% greater than the C_IN-MIN value. To determine the RMS input current rating (I_IN-RMS), use Equation 16.

Equation 16.

Because this approximation assumes there is no inductor ripple current, the value should be increased by 10-30% depending on the amount of ripple that is expected. Ceramic capacitors are the best choice for the input to the TPS92640 and TPS92641 devices due to their high ripple current rating, low ESR, low cost, and small size compared to other types. When selecting a ceramic capacitor, special attention must be paid to the operating conditions of the application. Ceramic capacitors can lose one-half or more of their capacitance at their rated DC voltage bias and also lose capacitance with extremes in temperature. Make sure to check any recommended deratings and also verify if there is any significant change in capacitance at the operating input voltage and the operating temperature.

8.1.5 NFETs

The TPS92640 and TPS92641 devices require two external NFETs for the switching regulator. The FETs should have a voltage rating at least 20% higher than the maximum input voltage to ensure safe operation during the ringing of the switch node. In practice, all switching converters have some ringing at the switch node due to the diode parasitic capacitance and the lead inductance. The NFETs should also have a current rating at least 50% higher than the average transistor current. Once NFETs are chosen, the power rating is verified by calculating the power loss.

8.2.1 TPS92640: Design Procedure

Figure 23. TPS92640 Design Procedure Schematic

8.2.1.2 Detailed Design Procedure

8.2.1.2.1 Set Output Voltage Feedback Ratio

For the desired output (V_OUT), R_VOUT1 and R_VOUT2 is calculated first with the desired feedback voltage, V_VOUT at approximately 2.5 V:

Equation 17.

8.2.1.2.2 Set Switching Frequency

The switching frequency is set as follows:

Equation 18.

8.2.1.2.3 Set Average LED Current

The average LED current (I_LED) is set by:

Equation 19.

8.2.1.2.4 Set Inductor Ripple Current

First, the expected duty cycle, D must be determined:

Equation 20.

With the inductor ripple current, Δi_L-PP specified and the expected duty cycle, the inductance (L) can be chosen:

Equation 21.

8.2.1.2.5 Set LED Ripple Current and Determine Output Capacitance, C_OUT

The LED ripple current (Δi_LED-PP ) is specified. With the target ripple current determined, the output capacitance (C_OUT) can be chosen using Equation 22.

Equation 22.

8.2.1.2.6 Choose N-Channel MOSFETs

The suggested minimum voltage rating, V_T-MAX and current rating, I_T-MAX are:

Equation 23.

Selecting a proper power MOSFET is critical in a power application, other than the SOA limits, the gate characteristic and the R_DSON can affect the system performance seriousely.

Also, the peak current limit (I_LIMIT) is governed by:

Equation 24.

Both the current limit threshold and MOSFET R_DSON are loosely specified and can vary a lot with temperature, input voltage and other operating conditions.

8.2.1.2.7 Choose Input Capacitance

Input capacitance is necessary to provide instantaneous current to the discontinuous portions of the circuit during the high side NFET on-time. The allowable input voltage ripple (Δv_IN-PP) is specified at approximately 3% Pk-Pk of V_IN. The minimum required capacitance (C_{IN_MIN}) to achieve this specification is:

Equation 25.

The necessary RMS input current rating (I_IN-RMS) can be approximated as follows:

Equation 26.

8.2.1.2.8 Set the Turnon Voltage and Undervoltage Lockout Hysteresis

With the desired turnon threshold voltage (V_{TURN_ON}) stated, the resistor divider network composing with R_UDIM1 and R_UDIM2 can be calculated with the equation in below.

Equation 27.

Then R_UDIM3 is optional and recommended for PWM. The R_UDIM3 can be calculated based on Equation 10 to provide the desired undervoltage lockout hysteresis (V_HYS).

8.2.2 TPS92640 – PWM Dimming Application

Figure 24. PWM Dimming Using UDIM Pin Schematic

8.2.2.2 Detailed Design Procedure

8.2.2.2.1 Calculate Operating Points

Calculate the operating points using Equation 28 to Equation 30, and assume approximately 90% conversion efficiency (η = 0.9).

Equation 28.

Equation 29.

Equation 30.

8.2.2.2.2 Output Voltage Feedback

Calculate the VOUT pin resistors by setting R_VOUT2 = 10 kΩ and calculating R_VOUT1.

Equation 31.

Choose R_VOUT1 = 120 kΩ.

8.2.2.2.3 Switching Frequency

Using the values calculated above choose a value of C_ON = 1 nF and calculate the value of R_ON:

Equation 32.

Choose the closest standard resistor value of R_ON = 26.1 kΩ.

8.2.2.2.4 Set the Feedback Reference and LED Current

To get a value of V_CS = 200 mV V_IADJ must be set to 2 V. Choose a value of R_IADJ1 = 10 kΩ and solve for R_IADJ2:

Equation 33.

Choose the standard resistor value of R_IADJ2 = 19.6 kΩ and solve for R_CS using Equation 34.

Equation 34.

R_CS = 0.2 Ω is a standard resistor value.

8.2.2.2.5 Calculate the Inductor Value

Because this is a PWM dimming application, TI does not recommend much output capacitance for faster current rise and fall times, so the inductor ripple current should be close to the 300-mA peak-to-peak LED ripple current. Calculate and inductor value that will give you 350-mA peak-to-peak inductor ripple current or less:

Equation 35.

Choose the standard value of L = 68 µH which results in an actual Δi_L-PP of 342 mA.

8.2.2.2.6 Calculate the Output Capacitor Value

Given the actual inductor ripple current of 342-mA peak-to-peak, use Equation 36 to calculate the required output capacitor value.

Equation 36.

Choose C_OUT = 0.1 µF.

8.2.2.2.7 Calculate the MOSFET Parameters

The MOSFETs must have a minimum voltage and current rating for the application. The minimum ratings are calculated using Equation 37 and Equation 38.

Equation 37.

Equation 38.

Choose MOSFETs that have a drain-to-source voltage rating of greater than 63 V and a current rating greater than 1.26 A.

8.2.2.2.8 Calculate the Minimum Input Capacitance

The minimum input capacitance to achieve 1.5-V peak-to-peak input voltage ripple is calculated using Equation 39.

Equation 39.

For PWM dimming applications more input voltage ripple will be present at the PWM dimming frequency. For these applications, TI recommends using 10 times the amount of minimum input capacitance or more. Choose C_IN = 10 µF.

8.2.2.2.9 Undervoltage Lockout and Hysteresis

Choose a value of R_UDIM1 = 100 kΩ and calculate the values of R_UDIM2 and R_UDIM3 using Equation 40 and Equation 41.

Equation 40.

Equation 41.

Choose the nearest standard resistor values of R_UDIM2 = 3.32 kΩ and R_UDIM3 = 19.1 kΩ.

8.2.2.3 Application Curve

Figure 25. UDIM Dimming Waveform