Table 2. Resistor Values for Noninverting Gains With 50-Ω Input/Output Match

Table 3. Resistor Values For Inverting Gains With 50-Ω Input/Output Match

9.1.3.1 Setting Resistor Values To Minimize Noise

The OPA659 provides a very low input noise voltage. To take full advantage of this low input noise, designers must pay careful attention to other possible noise contributors. Figure 38 shows the op amp noise analysis model with all the noise terms included. In this model, all the noise terms are taken to be noise voltage or current density terms in either nV/√Hz or pA/√Hz.

Figure 38. Op Amp Noise Analysis Model

The total output spot noise voltage can be computed as the square root of the squared contributing terms to the output noise voltage. This computation adds all the contributing noise powers at the output by superposition, then takes the square root to arrive at a spot noise voltage. Equation 1 shows the general form for this output noise voltage using the terms shown in Figure 38.

Equation 1.

Dividing this expression by the noise gain (G_N = 1 + R_F/R_G) gives the equivalent input-referred spot noise voltage at the noninverting input, as Equation 2 shows.

Equation 2.

space

space

Putting high resistor values into Equation 2 can quickly dominate the total equivalent input-referred noise. A source impedance on the noninverting input of 5 kΩ adds a Johnson voltage noise term equal to that of the amplifier alone (8.9 nV/Hz). While the JFET input of the OPA659 is ideal for high source impedance applications in the noninverting configuration of Figure 35 or Figure 36, both the overall bandwidth and noise are limited by high source impedances.

9.1.3.2 Frequency Response Control

Voltage-feedback op amps such as the OPA659 exhibit decreasing signal bandwidth as the signal gain increases. In theory, this relationship is described by the gain bandwidth product (GBP) shown in the Electrical Characteristics. Ideally, dividing the GBP by the noninverting signal gain (also called the Noise Gain, or NG) can predict the closed-loop bandwidth. In practice, this guideline is valid only when the phase margin approaches 90 degrees, as it does in high gain configurations. At low gains (with increased feedback factors), most high-speed amplifiers exhibit a more complex response with lower phase margins. The OPA659 is compensated to give a maximally-flat frequency response at a noninverting gain of +1 (see Figure 35). This compensation results in a typical gain of +1 bandwidth of 650 MHz, far exceeding that predicted by dividing the 350-MHz GBP by 1. Increasing the gain causes the phase margin to approach 90° and the bandwidth to more closely approach the predicted value of (GBP/NG). At a gain of +10, the OPA659 shows the 35-MHz bandwidth predicted using the simple formula and the typical GBP of 350 MHz. Unity-gain stable op amps such as the OPA659 can also be band-limited in gains other than +1 by placing a capacitor across the feedback resistor. For the noninverting configuration of Figure 36, a capacitor across the feedback resistor decreases the gain with frequency down to a gain of +1. For instance, to band-limit a gain of +2 design to 20 MHz, a 32-pF capacitor can be placed in parallel with the 249-Ω feedback resistor. This configuration, however, only decreases the gain from 2 to 1. Using a feedback capacitor to limit the signal bandwidth is more effective in the inverting configuration of Figure 37. Adding that same capacitance to the feedback of Figure 37 sets a pole in the signal frequency response at 20 MHz, but in this case it continues to attenuate the signal gain to less than 1. Note, however, that the noise gain of the circuit is only reduced to a gain of 1 with the addition of the feedback capacitor.

9.1.3.3 Driving Capacitive Loads

One of the most demanding, and yet very common, load conditions for an op amp is capacitive loading. The OPA659 is very robust, but care should be taken with light loading scenarios so that output capacitance does not decrease stability and increase closed-loop frequency response peaking when a capacitive load is placed directly on the output pin. When the amplifier open-loop output resistance is considered, this capacitive load introduces an additional pole in the signal path that can decrease the phase margin. Several external solutions to this problem have been suggested. When the primary considerations are frequency response flatness, pulse response fidelity, and/or distortion, the simplest and most effective solution is to isolate the capacitive load from the feedback loop by inserting a series isolation resistor, R_ISO, between the amplifier output and the capacitive load. In effect, this resistor isolates the phase shift from the loop gain of the amplifier, thus increasing the phase margin and improving stability. The Typical Characteristics show the recommended R_ISO versus capacitive load and the resulting frequency response with a 1-kΩ load (see Figure 24). Note that larger R_ISO values are required for lower capacitive loading. In this case, a design target of a maximally-flat frequency response was used. Lower values of R_ISO may be used if some peaking can be tolerated. Also, operating at higher gains (instead of the +1 gain used in the Typical Characteristics) requires lower values of R_ISO for a minimally-peaked frequency response. Parasitic capacitive loads greater than 2pF can begin to degrade the performance of the OPA659. Moreover, long PCB traces, unmatched cables, and connections to multiple devices can easily cause this value to be exceeded. Always consider this effect carefully, and add the recommended series resistor as close as possible to the OPA659 output pin (see the Layout section).

With heavier loads (for example, the 100-Ω load presented in the test circuits and used for testing typical characteristic performance), the OPA659 is very robust; R_ISO can be as low as 10 Ω with capacitive loads less than 5 pF and continue to show a flat frequency response.

space

space

9.1.3.4 Distortion Performance

The OPA659 is capable of delivering a low distortion signal at high frequencies over a wide range of gains. The distortion plots in the Typical Characteristics show the typical distortion under a wide variety of conditions. Generally, until the fundamental signal reaches very high frequencies or powers, the second harmonic dominates the distortion with a negligible third harmonic component. Focusing then on the second harmonic, increasing the load impedance improves distortion directly. Remember that the total load includes the feedback network: in the noninverting configuration, this network is the sum of R_F + R_G, while in the inverting configuration the network is only R_F (see Figure 36). Increasing the output voltage swing directly increases harmonic distortion. A 6dB increase in output swing generally increases the second harmonic by 12 dB and the third harmonic by 18 dB. Increasing the signal gain also increases the second-harmonic distortion. Again, a 6-dB increase in gain increases the second and third harmonics by about 6 dB, even with a constant output power and frequency. Finally, the distortion increases as the fundamental frequency increases because of the rolloff in the loop gain with frequency. Conversely, the distortion improves going to lower frequencies, down to the dominant open-loop pole at approximately 300 kHz.

Note that power-supply decoupling is critical for harmonic distortion performance. In particular, for optimal second-harmonic performance, the power-supply high-frequency 0.1-μF decoupling capacitors to the positive and negative supply pins should be brought to a single point ground located away from the input pins.

The OPA659 has an extremely low third-order harmonic distortion. This characteristic also shows up in the two-tone, third-order intermodulation spurious (IMD3) response curves (see Figure 19). The third-order spurious levels are extremely low (less than –100 dBc) at low output power levels and frequencies below 10 MHz. The output stage continues to hold these levels low even as the fundamental power reaches higher levels. As with most op amps, the spurious intermodulation powers do not increase as predicted by a traditional intercept model. As the fundamental power level increases, the dynamic range does not decrease significantly. For two tones centered at 10 MHz, with –2 dBm/tone into a matched 50-Ω load (that is, 0.5 V_PP for each tone at the load, which requires 2 V_PP for the overall two-tone envelope at the output pin), the Typical Characteristics show a 96-dBc difference between the test tones and the third-order intermodulation spurious levels. This exceptional performance improves further when operating at lower frequencies and/or higher load impedances.

Table 4. Design Parameters

Table 5. OPA659 TIA Component Values and Bandwidth for Various Diode Capacitance and Gains